Wireless communications devices include two essential components—a transmitter and a receiver. In mobile wireless communications devices, such as cellular handsets, the transmitter and receiver are combined into a single unit and share a common antenna. When combined in this manner, the transmitter and receiver are collectively referred to as a “transceiver.”
Transceivers are generally categorized as being either “half-duplex” or “full-duplex”. In a half-duplex transceiver, only one of the transmitter and receiver is permitted to operate at any give time. In a full-duplex transceiver the transmitter and receiver operate simultaneously. To avoid interference between transmitted and received signals in full-duplex operation, the transmitter and receiver are designed to transmit and receive in different and ideally non-overlapping frequency bands. Whether half-duplex or full-duplex operation is used is usually determined by the wireless technology involved. For example, second generation (2G) cellular technologies such as Global System for Mobile communications (GSM) and Enhanced Data rates for GSM Evolution (EDGE) employ half-duplex operation, while third generation (3G) cellular technology based on the Wideband Code Division Multiple Access (W-CDMA) standard employs full-duplex operation.
FIG. 1 is a simplified drawing of a full-duplex transceiver 100. The full-duplex transceiver 100 includes a transmitter 102, a receiver 104, an antenna 106, and a duplexer 108 that couples the transmitter 102 and receiver 104 to the antenna 106. The transmitter 102 includes an upconverter 110 and a power amplifier (PA) 112, which operate to upconvert and amplify in-phase (I) and quadrature phase (Q) baseband signals to radio frequency (RF) signals centered in a transmit (Tx) band. The Tx signals are passed through the duplexer 108 and fed to the antenna 106, which radiates the Tx signals over the air to a remote receiver. The receiver 104 includes a low-noise amplifier (LNA) 114 and a downconverter 116. The LNA 114 amplifies receive (Rx) signals received in a Rx band, and the downconverter 116 downconverts the amplified Rx signals from RF to baseband. The duplexer 108 provides a filtering and isolation function, which ideally prevents Tx signals from the transmitter 102 from leaking into and saturating the front-end of the receiver 104.
One of the most difficult challenges in the design of the full-duplex transceiver 100 involves the design of the transmitter 102. Not only should the transmitter 102 be power efficient in order to conserve battery life, it should also transmit only within a designated Tx band in order to avoid desensitizing the receiver 104. Unfortunately, a transmitter having both of these desirable attributes is difficult to design.
Conventional transmitter architectures are based on what is known as a quadrature modulator. FIG. 2 is a block diagram of quadrature-modulator-based transmitter 200. The quadrature-modulator-based transmitter 200 comprises a quadrature modulator 202 including an I-channel mixer 210, Q-channel mixer 212, local oscillator (LO) 214, 90° phase shifter 216, and summer 218; a surface acoustic wave filter (SAW) 204; a power amplifier (PA) 206; and an antenna 208.
The quadrature modulator 202 operates to upconvert information to be transmitted and contained in I and Q baseband signals to RF by modulating the information onto orthogonal RF carrier signals generated by the LO 214 and the 90° phase shifter 216. The summer 218 combines the upconverted signals and couples the summed result to an input of the SAW filter 204, which operates as a bandpass filter. Finally, the bandpass-filtered signal from the SAW filter 204 is amplified by the PA 206 and radiated over the air by the antenna 208 to a remote access point or cellular base station.
One desirable characteristic of the quadrature-modulator-based transmitter 200 is that the frequency and phase of the RF carrier signal can be modulated simply by manipulating the amplitudes of the I and Q baseband signals. However, a drawback is that it is not very power efficient. In an effort to increase spectral efficiency, many state-of-the-art communications systems employ nonconstant-envelope signals. To prevent clipping of the signal peaks of these nonconstant-envelope signals in the quadrature-modulator-based transmitter 200, the signal levels must be reduced before being introduced to the input of the PA 206, and the PA 206 must be configured to operate in its linear region of operation. Unfortunately, linear PAs configured to operate at reduced drive levels are not very power efficient. This lack of power efficiency is a major concern, particularly in battery-powered applications.
The linearity versus efficiency trade-off of the quadrature-modulator-based transmitter 200 can be avoided by using an alternative type of transmitter known as a polar modulation transmitter. As explained below, a polar modulation transmitter temporarily removes the amplitude information from nonconstant-envelope signals so that the polar modulation transmitter's PA can be configured to operate in its nonlinear region, where it is much more efficient at converting power from the transmitter's power supply into RF power.
FIG. 3 is a simplified drawing of a polar modulation transmitter 300. The polar modulation transmitter 300 comprises a rectangular-to-polar converter (or Coordinate Rotation Digital Computer (CORDIC)) converter 302; an amplitude modulator 304 configured in an amplitude path; a phase modulator 306 configured in a phase path; a PA 308; and an antenna 310.
The CORDIC converter 302 operates to convert rectangular-coordinate I and Q baseband signals into polar-coordinate amplitude and phase component signals Am(t) and Pm(t), according to the mapping functions: Am(t)=√{square root over (I(t)2+Q(t)2)}{square root over (I(t)2+Q(t)2)} and Pm(t)=tan−1[Q(t)/I(t)]. The amplitude modulator 304 modulates a direct current (DC) power supply Vsupply (e.g., as supplied from the communication device's battery) according to amplitude variations represented in the amplitude component signal Am(t). The resulting amplitude modulated power supply signal Vs(t) is coupled to the power supply port of the PA 308. Meanwhile, the phase modulator 306 operates to modulate an RF carrier signal with phase information represented in the phase component signal Pm(t). Because the resulting phase-modulated RF carrier signal RFin has a constant envelope, the PA 308 can be configured to operate in its nonlinear region of operation, where it is efficient at converting power from the DC power supply Vsupply to RF power. Typically, the PA 308 is implemented as a Class D, E or F switch-mode PA 308 operating in compression, so that the output power of the PA 308 is directly controlled by the amplitude modulated power supply signal Vs(t) applied to the power supply port of the PA 308. Effectively, the PA 308 operates as a multiplier, amplifying the constant-envelope phase-modulated RF carrier signal according to amplitude variations in the amplitude modulated power supply signal Vs(t) to produce the desired amplitude- and phase-modulated RF carrier signal RFout.
Although the polar modulation transmitter 300 is more power efficient than the quadrature-modulator-based transmitter 200, it has a number of drawbacks of its own. On problem relates to the fact that the bandwidths of the amplitude and phase component signals Am(t) and Pm(t) are typically wider than the bandwidths of the rectangular-coordinate I and Q signals. This so-called “bandwidth expansion” occurs due to the fact that the mapping functions: Am(t)=√{square root over (I(t)2+Q(t)2)}{square root over (I(t)2+Q(t)2)} and Pm(t)=tan−1[Q(t)/I(t)] involve a nonlinear process that tends to spread the power spectra of the resulting modulation signals over a frequency range beyond the boundaries of the desired Tx band.
The level of bandwidth expansion that occurs depends in large part on the modulation format used. As mentioned above, many existing technologies such as orthogonal frequency division multiplexing (OFDM), and other existing or soon-to-be deployed cellular technologies, such as W-CDMA, High-Speed Packet Access (HSPA) and Long Term Evolution (LTE) technologies, employ nonconstant-envelope modulation formats, which produce signal trajectories passing through (or very close to) the origin. As the signal traverses through or near the origin, the magnitude of the amplitude component signal Am(t) rapidly decreases and then increases, while the phase of the phase component signal Pm(t) experiences a rapid phase jump of + or −180°.
A high level of bandwidth expansion results in a phase component single signal Pm(t) having a bandwidth that is difficult to translate to RF since the phase modulator 306 is usually capable of providing a linear response only over a narrowly-defined frequency range. Bandwidth expansion in the amplitude component signal Am(t) can also be problematic. To maximize efficiency, the amplitude modulator 304 is typically implemented as a switch-mode power supply. However, when the bandwidth of the amplitude component signal Am(t) is wider than the switching capabilities of the switch-mode power supply's switching transistors, the switch-mode converter is unable to accurately track the envelope of the amplitude component signal Am(t) and out-of-band signal energy is generated.
Another problem with the polar modulation transmitter 300 relates to the difficulty the PA 308 has in generating low output powers without also producing significant amounts of out-of-band distortion. Some wireless communications standards require that the transmitter of a communications device be capable of controlling its output power over a wide dynamic range. For example, the W-CDMA standard requires a transmitter of a cellular handset to be capable of controlling its average output power over a 70 dB range, including a lower limit of around −50 dBm. Unfortunately, this range of power control, particularly at low average output powers, is difficult to achieve in the polar modulation transmitter 300. In the polar modulation transmitter 300, average RF output power is controlled by the PA 308 in response to the magnitude of the amplitude modulated power supply signal Vs(t) supplied to the power supply port of the PA 308. When the magnitude of the modulated power supply signal Vs(t) is high and the PA 308 is operating in compression, the average RF output power of the PA 308 is proportional to the square of the amplitude modulated power supply signal Vs(t), and the amplitude- and phase-modulated RF carrier signal RFout responds fairly linearly to changes in the amplitude modulated power supply signal Vs(t). However, as illustrated in FIG. 4, when the magnitude of the modulated power supply signal Vs(t) is lowered to achieve lower RF output powers, these relationships no longer hold and the PA 308 responds in a highly nonlinear fashion. The nonlinear response of the PA 308 at low output powers can lead to significant out-of band distortion, beyond that due to the effects of bandwidth expansion alone.
When the polar modulation transmitter 300 is used to implement the transmitter 102 of the full-duplex transceiver 100, the out-of-band Tx signal energy generated from the effects of bandwidth expansion and the nonlinear characteristics of the PA 308 at low output powers can hinder the performance of the receiver 104. As illustrated in FIG. 5, due to finite Tx-Rx isolation provided by the duplexer 108, the out-of-band Tx signal energy leaks through the duplexer 108 and into the front-end of the receiver 104, where it appears as receive band noise. Receive band noise is of particular concern when the magnitude of the leaked signal energy is large compared to the magnitude of the desired Rx signals. It is also highly undesirable, since it desensitizes the receiver 104 and makes it difficult, or in some cases even impossible, to comply with out-of-band noise limitation standards.